This invention relates to a frequency-division multiplexing transceiver apparatus, wave-number-division multiplexing transceiver apparatus and method thereof, and more particularly to a frequency-division multiplexing transceiver apparatus, wave-number-division multiplexing transceiver apparatus and method for transmitting data in a frequency spectrum or wave-number spectrum.
DS-CDMA (Direct Sequence-Code Division Multiple Access) multiplies a narrow-band transmission signal by a spreading code in order to spread and transmit that transmission signal over a wider band. In DS-CDMA, when each of a plurality of mobile stations sends a transmission signal upon multiplying it by a spreading code having a certain spreading factor SF, the information transmission speed becomes 1/SF. Therefore, in order to achieve a frequency utilization efficiency that is equivalent to that of TDMA, it is necessary in DS-CDMA to accommodate a number of signals that is equal to SF number of mobile stations. However, in an actual wireless propagation environment on the uplink, the effect of Multiple Access Interference (MAI), in which the signals from each of the mobile stations interfere with each other, becomes dominant due to differences in propagation conditions from each mobile station to the base station, for example, due to differences in propagation-delay time or propagation-path fluctuation, and thus the rate of frequency utilization decreases.
Therefore, IFDMA (Interleaved Frequency Division Multiple Access) is being studied as a wireless modulation method that is capable of reducing the effects of MAI in next-generation mobile communications (see the specification of JP2004-297756A, and Goto, et al., “Investigations on Packet Error Rate of Variable Spreading and Chip Repetition Factors (VSCRF)-CDMA Wireless Access in Reverse Link Multi-cell Environment”, The Institute of Electronics, Information and Communication Engineers, Technical Report of IEICE, RCS2004-84 (204-206). This IFDMA modulation method transmits a transmission signal upon multiplying the signal by a phase that changes at a speed specific to the mobile station, thereby reducing MAI by placing the signals from each of the mobile stations on a frequency axis in such a manner that the signals will not overlap each other on the frequency axis.
FIG. 21 is a block diagram showing the structure of a mobile station that uses an IFDMA modulation method, and FIG. 22 is a drawing that explains an IFDMA symbol. A channel encoder 1a performs channel encoding by applying error-correction encoding such as turbo encoding or convolutional encoding to an entered binary information sequence, and a data modulator 1b converts the channel-encoded data to I, Q complex components (symbols) in QPSK, for example. A symbol transmitted in one frame of IFDMA is referred to as an “IFDMA symbol”, and one IFDMA symbol is composed of Q-number of symbols S0, S1, S2, S3 as shown in (a) of FIG. 22 (Q=4 in the figure).
A symbol-repetition-and-rearrangement unit 1c compresses the time domains of the four symbols S0, S1, S2 and S3 of the IFDMA symbol, and repeatedly generates each symbol L times (L=4 in the figure), as well as rearranges the repeatedly generated symbols and places them in the same arrangement as that of the symbol sequence S0, S1, S2, S3 (see (b) of FIG. 22). By taking Tc to be the sample period, the period Ts of symbol repetition will satisfy the relation Ts=Tc×Q. A phase-rotation unit 1d has a complex multiplier CML that performs mobile-station specific phase rotation for each symbol in the repetitive symbol sequence (see (c) of FIG. 22), and a wireless transmitter 1e performs up-conversion of the signal that is input from the phase-rotation unit 1d from baseband frequency to radio frequency, after which it amplifies the signal and transmits it from an antenna.
When the time domains of the transmission-symbol sequence S0, S1, S2, S3 are compressed and each transmission symbol is repeatedly generated a prescribed number of times (L times), and each of the symbols of the repetitive-symbol sequence are rearranged so as to have the same arrangement as that of the symbol sequence S0, S1, S2, S3, the repetitive-symbol sequence after rearrangement will have a comb-tooth-shaped frequency spectrum as shown in (a) of FIG. 23. Also, by performing phase rotation that varies at a speed that is specific to the mobile station for each of the symbols of the rearranged repetitive-symbol sequence, the spectral positions of the comb-tooth-shaped frequency spectrum shift as shown in (a) to (d) of FIG. 23, and frequency-division multiplex transmission becomes possible. In other words, when the speed of phase rotation is zero, the frequency spectrum of the output signal from the phase-rotation unit 1d will have comb-tooth-shaped frequency spectrum characteristics as shown in (a) of FIG. 23, and as the amount of change in the phase rotation per unit time Tc increases, the frequency spectrum will shift as shown in (a) to (d) of FIG. 23.
An NCO (Numerically Controlled Oscillator) 1g calculates the amount of phase rotation θ per unit time Tc, and the complex multiplier of the phase-rotation unit 1d performs phase rotation specific to the mobile station for each symbol of the repetitive-symbol sequence and executes frequency shift processing.
In a case where Q-number of symbols are repeated L times, the phase θk(t) that is output from the NCO 1g is given by the following equation:
                                                        θ              k                        ⁡                          (              t              )                                =                                                    k                ·                2                            ⁢              π              ⁢                                                          ⁢                                                W                  L                                ·                t                                      =                                          k                ·                2                            ⁢              π              ⁢                                                          ⁢                                                1                                      L                    ·                    Q                    ·                    Tc                                                  ·                t                                                    ⁢                                  ⁢                  W          =                      1            Ts                          ⁢                                  ⁢                                  ⁢                  QW          =                      1            Tc                                              (        1        )            where W is the symbol frequency, and k is a value that corresponds to the mobile station and is any one value among 0, 1, 2, . . . L−1. NCO 1g outputs the phase θk(t), which has been calculated according to Equation (1), at the period Tc, and is so adapted that the amount of phase rotation will be 2π at the IFDMA period (=L·Q·Tc=16Tc) (such that the phase will make one full cycle).
In NCO 1g, a frequency-shift-setting unit 1h sets the amount Δω of change of phase rotation (angular speed) per unit time Tc. That is, using the parameters k, L and Q, the unit 1h calculates and outputs the angular speed Δω according to the following equations:
                              Δω          =                                                    k                ·                2                            ⁢              π              ⁢                                                          ⁢                              W                L                                      =                                          k                ·                2                            ⁢              π              ⁢                                                          ⁢                              1                                  L                  ·                  Q                                                                    ⁢                                  ⁢                  f          =                                    Δω                              2                ⁢                                  π                  ·                  Tc                                                      =                          k                              L                ·                Q                ·                Tc                                                                        (        2        )            A rotation-phase-amount-setting unit 1i comprises an adder ADD and a delay unit DLY for applying a delay time T (=Tc), and performs a calculation according to the following equation every unit time Tc to increase the rotation phase θ by Δω at a time and outputs the result.θ=θ+Δω  (2a)A converter 1j calculates I, Q components (x, y) in a complex plane of the rotation phase amount θ and inputs these components to the phase-rotation unit 1d. By taking the symbols of the repetitive-symbol sequence to be S (=X+jY), the phase-rotation unit 1d performs a calculation according to the following equation and outputs the calculation result.(X+jY)·(x+jy)In actuality, the complex multiplier CML of the phase-rotation unit 1d calculates and outputs (Xx−Yy) and (Xy+Yx) for each real-number and imaginary-number part.
If k=0, the amount of phase shift will be zero (Δω=0, and the frequency spectrum will become as shown in (a) of FIG. 23. If k=1, the amount of phase shift will become Δω=2π/L×Q according to Equation (2), and if Q=L=4, then the phase will change in increments of π/8 and the frequency spectrum will become as shown in (b) of FIG. 23. Also, if k=2, the amount of phase shift will become Δω=4π/L×Q according to Equation (2). If Q=L=4, then the phase will change in increments of 2π/8 for each Tc, and the frequency spectrum will become as shown in (c) of FIG. 23. Moreover, if k=3, then the amount of phase shift will become Δω=6π/L×Q according to Equation (2). If Q=L=4, then the phase will change in increments of 3π/8 for each Tc, and the frequency spectrum will become as shown in (d) of FIG. 23. As a result, even when a plurality of mobile stations access the same base station simultaneously, the frequency spectrum of each mobile station will be orthogonal on the frequency axis, and it is possible to reduce interference among transmission signals.
In mobile wireless communication, depending on the propagation path, MPI (Multi-Path Interference) occurs, and line quality becomes poor. Therefore, in prior IFDMA, in order to reduce MPI as explained in paragraphs [0010] to [0014] of JP2004-297756A, a multi-path interference canceller is used. However, in the method of using this multi-path interference canceller, the amount of processing necessary increases, and there is a problem with traceability.
Therefore, an OFDM method that is capable of lowering the effect of MPI is being studied as a modulation method. However, in the OFDM method, transmission symbols are multiplexed on orthogonal frequencies, so the Peak to Average Power Ratio (PAPR) becomes large and the transmission efficiency of the transmission amplifier becomes poor. In order to prevent the PAPR from increasing, a single portion that is greater than a threshold value is deleted by a clipping process in the transmission unit in order to suppress peak power, and this makes it possible to reduce the peak power that is input to the transmission amplifier. However, there is a problem in that there is an increase in the rate of code errors due to suppression of the peak power.